Receiver circuit having an optical reception device

ABSTRACT

A receiver circuit having an optical reception device and having an amplifier connected to the reception device, the amplifier also having a circuit for setting the operating point of the amplifier and also at least one control terminal of the circuit, by which the operating point of the amplifier can be selectively changed between at least two values at the user end. The receiver circuit according to the invention enables a noise optimization of the amplifier by virtue of an adjustability of the operating point of the amplifier.

RELATED APPLICATION

The present application claims priority of U.S. Patent Application Ser.No. 60/540,870 filed by Karl Schrodinger on Jan. 30, 2004.

FIELD OF THE INVENTION

The invention relates to a receiver circuit having an optical receptiondevice and having an amplifier connected downstream of the opticalreception device. In particular, the invention relates to a receivercircuit having a transimpedance amplifier for optical transmissionsystems.

BACKGROUND OF THE INVENTION

Receiver circuit having an optical reception device are known in whichlight incident on the optical reception device—for example, light froman optical waveguide of an optical data transmission system—is detectedby the optical reception device with formation of an electrical signal(e.g. a photocurrent) and the electrical signal is subsequentlyamplified by the amplifier connected downstream.

An optical receiver circuit having an optical reception device andhaving an amplifier connected downstream is described for example in thearticle “High Gain Transimpedance Amplifier in InP-Based HBT Technologyfor the Receiver in 40-Gb/s Optical-Fiber TDM Links” (Jens Müllrich,Herbert Thurner, Ernst Müllner, Joseph F. Jensen, Senior Member, IEEE,William E. Stanchina, Member, IEEE, M. Kardos, and Hans-Martin Rein,Senior Member, IEEE—IEEE Journal of Solid State Circuits, vol. 35, No.9, September 2000, pages 1260 to 1265). In the case of this receivercircuit, at the input end there is a differentially operatedtransimpedance amplifier—that is to say a differentialamplifier—connected by one input to a photodiode as reception device.The other input of the differentially operated transimpedance amplifieris connected to a DC amplifier which feeds a “correction current” intothe differential amplifier for the purpose of offset correction of thephotocurrent of the photodiode. The magnitude of this “correctioncurrent” that is fed in amounts to half the current swing of thephotodiode during operation.

An optical receiver circuit is always subject to noise. In the case ofan optical receiver circuit having a transimpedance amplifier, the mostimportant noise sources are the input transistor of the transimpedanceamplifier and the transimpedance impedance.

There is a need for receiver circuits which have a favorable noisebehavior.

SUMMARY OF THE INVENTION

The invention provides a receiver circuit, which has: an opticalreception device and an amplifier connected to the reception device, theamplifier having a circuit for setting the operating point of theamplifier and also at least one control terminal of the circuit, bymeans of which the operating point of the amplifier can be changed overbetween at least two values at the user end (i.e., the operating pointis user configurable).

The present invention is based on the concept of providing an amplifieroperating point that can be selectively changed over for the purpose ofnoise optimization. In this case, it is preferably the operating pointof an input transistor of the amplifier that is set, the noise of whichpredominates over the noise of the amplifier. In this case, the noise ofthe input transistor can be set by way of the operating point thereof.

In a preferred refinement, the circuit for setting the operating pointof the amplifier forms a setting of the operating point of the inputtransistor by setting the current in the input transistor. In this case,the circuit for setting the operating point of the amplifier ispreferably formed between the input transistor and a reference point, atwhich the operating voltage is present. Changing over the current in theinput transistor changes over the operating point thereof. This isaccompanied by an altered noise, in which case it holds true that thenoise in the input transistor likewise decreases as the current in thetransistor decreases.

Preferably, the circuit for setting the operating point of the amplifieris formed by an impedance network with at least one switching device,which can be changed over at the user end by means of the at least onecontrol terminal, the total impedance of the impedance network beingaltered. In particular, it is preferably provided that the impedancenetwork has a plurality of ohmic resistors, which can be connected inand disconnected by means of the at least one switching device and theat least one control terminal.

In one development of the invention, the amplifier furthermore has atleast one gain control terminal, by means of which the gain of theamplifier can be changed over at least between two gain values at theuser end. This enables an optimal optical sensitivity: this is becausethe adjustability of the gain of the amplifier makes it possible to setthe maximum gain of the amplifier depending on the prescribed bandwidth,or bandwidth to be achieved, of the receiver circuit. By way of example,on account of the approximately constant bandwidth (B)-gain (V) product(B*V=K; K results from the individual configuration of the receivercircuit), it is possible to set the maximum gain V and thus the maximumsensitivity of the receiver circuit by choosing V=K/B. The receivercircuit can thus be used optimally for different data rates. Thus, onaccount of the gain that can be changed over, the receiver circuit canbe individually adapted for example to transmission rates of 1 Gbps(gigabit per second), 2 Gbps or 4 Gbps.

A further essential advantage of the receiver circuit with a gain thatcan be changed over consists in its optimal noise behavior. By way ofexample, if a photodiode is used as the reception device and atransimpedance amplifier is used as the amplifier, then the currentnoise has a particularly relevant part to play in the amplifier.However, the current noise which is attributable to the transimpedanceamplifier generally becomes lower toward higher gains of the amplifier,so that, when the optimum—that is to say maximum—gain is chosen, thecurrent noise of the amplifier also decreases. However, with other typesof amplifier, too, it generally holds true that the signal-to-noiseratio becomes better in the case of a higher gain. In summary, the noisebehavior of the receiver circuit can be improved further as a result ofthe user-end setting of the optimum gain value depending on therespective bandwidth requirement.

The amplifier preferably has a feedback impedance, which influences thegain of the amplifier. The impedance of the feedback impedance can thenbe set externally at the user end by means of the at least one gaincontrol terminal. In particular, the resistance of the feedbackimpedance should be able to be set at the user end by means of the atleast one control terminal.

In order to be able to ensure the adjustability of the impedance of thefeedback impedance in a particularly simple manner, one advantageousdevelopment of the receiver circuit proposes that the feedback impedanceis formed by an impedance network with at least one switching device,which can be changed over at the user end by means of the at least onecontrol terminal and which alters the impedance or the resistance of theimpedance network in the case of a changeover. In this case, theswitching device is preferably formed by a switching transistor, inparticular a MOS-FET transistor.

Another advantageous development of the receiver circuit proposes thatthe feedback impedance is formed by an impedance network with at leastone variable impedance, the impedance of which can be set at the userend within a predetermined impedance range at least approximatelylinearly by means of the control terminal. The variable impedance may beformed for example by a transistor, in particular a MOS-FET transistor.

In transimpedance amplifiers, the bandwidth is approximatelyproportional to the reciprocal of the feedback impedance, that is to sayto 1/feedback impedance, since the gain is proportional to the feedbackimpedance. In this case, the gain is determined by the so-calledtransimpedance (=output voltage/input current).

BRIEF DESCRIPTION OF THE DRAWINGS

The invention is explained in more detail below using an exemplaryembodiment with reference to the figures, in which:

FIG. 1 shows an exemplary embodiment of a receiver circuit having anoptical reception device and an amplifier connected to the receptiondevice, said amplifier having a feedback impedance;

FIG. 2 shows an exemplary embodiment of the feedback impedance of theamplifier of the receiver circuit of FIG. 1;

FIG. 3 shows an exemplary embodiment of the amplifier of the receivercircuit of FIG. 1; and

FIG. 4 shows the spectral noise power of the amplifier of FIG. 3 forthree different circuit states of the amplifier.

DESCRIPTION OF A PREFERRED EXEMPLARY EMBODIMENT

FIG. 1 reveals a receiver circuit 10 with a photodiode 20 as opticalreception device. A transimpedance amplifier 30 is arranged downstreamof the photodiode 20. The transimpedance amplifier 30 comprises avoltage amplifier 40, for example an operational amplifier, and afeedback impedance 50. The feedback impedance 50 is connected to theinput end of the operational amplifier 40 by its terminal E50 and to theoutput end of the operational amplifier 40 by its terminal A50.

At the output end, the transimpedance amplifier 30 is additionallyconnected to a differential amplifier 60, which amplifies the outputsignal Sa of the transimpedance amplifier 30. A further amplification ofthe signal is effected by a second differential amplifier 70 arrangeddownstream of the first differential amplifier 60.

FIG. 1 furthermore reveals a control circuit 80, which, at the inputend, is connected to the two outputs A70 a and A70 b of the seconddifferential amplifier 70. The control circuit 80 additionally has acontrol input S80, via which a user-end control signal Sb can be fedinto the control circuit 80. The control input S80 thus forms a controlterminal S10 of the receiver circuit 10.

By an output A80, the control circuit 80 is connected to a controlterminal S30 of the transimpedance amplifier 30 and thus to a controlinput S50 of the feedback impedance 50. Via said control input S50, thecontrol circuit 80 can define the impedance, in particular also theresistance, of the feedback impedance 50 by means of an impedancespecification signal Sr formed from the user-end control signal Sb.

Furthermore, the optical receiver circuit is equipped with a DCC circuit90 (DCC: Duty Cycle Control), which effects a control of the opticalreceiver circuit. The DCC circuit 90 or the duty cycle control (offsetcontrol) formed by it controls the sampling threshold for the downstreamdifferential amplifiers, so that the signal is sampled at the 50% valueof the amplitude and, as a result, no signal pulse distortions (dutycycle) are produced. This can be effected by feeding a current into arespective one of the preamplifiers (transimpedance amplifiers) or elseby feeding in a voltage at the inputs of the differential amplifiersdirectly.

The photodiode 20 is connected via a low-pass filter 100 formed from acapacitor C_(PD) and a resistor R_(PD), a supply voltage VCC1 beingapplied to said filter. The low-pass filter 100 serves to “filter out”possible interference signals on the supply voltage VCC.

The optical receiver circuit 10 in accordance with FIG. 1 is operated asfollows:

When light is incident, a photocurrent I_(photo) is generated by thephotodiode 20 and fed into the transimpedance amplifier 30, where thephotocurrent is amplified to form the output signal Sa. The electricaloutput signal Sa is amplified further by the two differential amplifiers60 and 70 to form an amplified output signal Sa′ and passes to theoutput A10 of the optical receiver circuit 10; the output A10 of theoptical receiver circuit 10 is thus formed by the two outputs A70 a andA70 b of the second differential amplifier 70.

The gain of the transimpedance amplifier 30 is set at the user end bymeans of the control signal Sb via the control terminal S80 of thecontrol circuit 80 or via the control terminal S10 of the receivercircuit 10. For this purpose, the control signal Sb generated at theuser end passes to the control circuit 80, which, with its impedancespecification signal Sr, sets the resistance of the feedback impedance50. This is because the magnitude of the resistance (|R|) of thefeedback impedance 50 directly influences the gain of the transimpedanceamplifier 30 because the following holds true:Sa=|R|*I _(photo)

Thus, in the case of the arrangement in accordance with FIG. 1, the gainof the transimpedance amplifier 30 can be prescribed at the user end bymeans of the control signal Sb.

When prescribing an optimum gain value for the transimpedance amplifier30, it is necessary to take account of the bandwidth B respectivelyrequired. In concrete terms, a very large gain is possible given a verysmall bandwidth, whereas only a very small gain can be achieved given avery large bandwidth. In concrete terms, this is due to the fact that,to a first approximation, the bandwidth-gain product (V*B) of thereceiver circuit 10 is approximately constant and is prescribed by theindividual configuration of the receiver circuit. The product V*B can bedetermined by measurement, for example.

Thus, if a specific bandwidth is prescribed or is at least to beachieved, then the maximum permissible gain can be derived from this atthe user end. A corresponding gain value is then set by the controlcircuit 80 through the selection of the corresponding magnitude of thefeedback impedance 50.

The desired gain can therefore be prescribed at the user end via thecontrol input S80 and thus by means of the control signal Sb. As analternative—given a corresponding configuration of the control circuit80—a bandwidth to be achieved can also be communicated to the controlcircuit 80 at the user end by means of the control signal Sb, from whichthe maximum permissible gain V is then determined by the control circuit80 in accordance with the mathematical relationship mentioned above andis communicated to the transimpedance amplifier 30 via the output A80and the control terminal S50.

In connection with FIG. 1, the user-end control signal Sb was conductedto the transimpedance amplifier 30 via the control device 80. Instead ofthis, the user-end control signal Sb may also be applied directly to thecontrol terminal S30 of the transimpedance amplifier 30.

Moreover, the transimpedance amplifier 30, the two differentialamplifiers 60 and 70, the control circuit 80 and the DCC circuit 90 mayalso be regarded as one “amplifier unit” or as one “amplifier” whosecontrol terminal for feeding in the user-end control signal Sb is formedby the terminal S80 of the control circuit 80.

FIG. 2 illustrates an exemplary embodiment of a feedback impedance 50 inaccordance with FIG. 1. The feedback impedance is formed by userconfigurable impedance network. The illustration reveals an ohmicresistor RF1, with which capacitors CF1, CF2, CF3, CFC1 and CFC2 areconnected in parallel. In addition, further ohmic resistors RF2 and RF3are connected in parallel with the resistor RF1.

As can be discerned in FIG. 2, the resistor RF2 and the capacitor CF2are connected in parallel and are connected to a switching transistor210. If the switching transistor 210 is switched off, then the resistorRF2 and the capacitor CF2 play no part in the total impedance of theimpedance network. By contrast, if the switching transistor 210 isswitched on, then the resistors RF1 and RF2 form an ohmic parallelconnection, with the result that the total resistance of the impedancenetwork is reduced. The capacitor CF2 correspondingly increases thetotal capacitance of the impedance network since the capacitor CF2 isadded to the capacitor CF1.

The resistor RF3 and the capacitor CF3 can be connected in parallel withthe first resistor RF1 in a corresponding manner by means of a secondswitching transistor 220.

FIG. 2 furthermore reveals a MOS-FET transistor 230, which represents alinearly controllable resistor. Depending on the gate voltage applied tothe MOS-FET transistor, a transistor resistor is produced which isconnected in parallel with the first resistor RF1 and thus linearlyreduces the total resistance of the impedance network. The resistance ofthe impedance network can be set in a continuously variable manner byapplication of the gate voltage.

Via a third switching transistor 240 and a fourth switching transistor250, the capacitor CFC1 and the capacitor CFC2 can likewise be connectedin parallel with the first resistor RF1, or else “disconnected”.

FIG. 2 furthermore reveals a coding device 300, the input E300 of whichforms the control terminal S50 of the feedback impedance 50 inaccordance with FIG. 1. At the output end, the coding device 300 isconnected to the four switching transistors 210, 220, 240 and 250 andalso to the linearly operating MOS-FET transistor 230.

The coding device 300 serves to recode the impedance specificationsignal Sr formed by the control circuit 80 in such a way that thefeedback impedance 50 or the impedance network forms the desiredimpedance and the transimpedance amplifier 30 thus achieves the requiredgain.

The impedance network is driven as follows for the operation of thereceiver circuit in accordance with FIG. 1:

The resistor RF1 serves for setting the largest gain and thus thesmallest bandwidth of the transimpedance amplifier 30. In this operatingmode—that is to say with the smallest bandwidth—the second resistor RF2and the third resistor RF3 are disconnected by the two switchingtransistors 210 and 220. The capacitor CF1 serves for compensationagainst oscillation tendencies of the receiver circuit 10.

If a higher data rate is required, then the second resistor RF2 isconnected in, by way of example; a lower transimpedance impedance isthus produced as a result of the two resistors RF1 and RF2 beingconnected in parallel, as a result of which the gain of thetransimpedance amplifier 30 is reduced and the bandwidth is increased.

As a result of further connection—for example of the third resistorRF3—the resistance of the feedback impedance 50 and thus the gain of thetransimpedance amplifier 30 can be reduced further, as a result of whichthe bandwidth is increased further. The compensation capacitors CF2 andCF3 that are necessary, if appropriate, for compensation againstoscillation tendencies are additionally connected in at the same time asthe two resistors RF2 and RF3 by the two switching transistors 210 and220. In this case, the transistors 210, 220, 230, 240 and 250 arechanged over by the control signal SV by means of the coding device 300.

The function of the MOS-FET transistor 230, which is likewise controlledby the coding device 300 and the control circuit 80, serves primarilyfor amplitude control. If the output power of the transimpedanceamplifier rises increasingly, then the transistor 230 is drivenlinearly, so that the feedback impedance (transimpedance impedance) 50of the transimpedance amplifier 30 is continuously decreased:overdriving of the transimpedance amplifier 30 can be prevented in thisway. In order to be able to identify an increase in the output power ofthe transimpedance amplifier 30, the control circuit 80 in accordancewith FIG. 1 is connected to the output signals Sa′ and −Sa′ of thefurther differential amplifier 70.

The additional capacitors CFC1 and CFC2 can be connected in with theassociated switching transistors 240 and 250 in order to avoidoscillations; this may be necessary particularly when the feedbackimpedance 50 of the transimpedance amplifier 30 is decreased linearly onaccount of the MOS-FET transistor 230.

In summary, in the case of the exemplary embodiment in accordance withFIG. 2, the feedback impedance 50 is reduced by resistors and/orcapacitors being connected in “parallel”. Instead of this or inaddition, a changeover of the impedance of the feedback impedance 50 mayalso be achieved through a series circuit of connectable resistorsand/or connectable capacitors.

The coding device 300 may be formed for example by an integrated circuitwhich correspondingly converts the impedance specification signal Sr insuch a way that the transistors 210, 220, 230, 240 and 250 are driven inthe manner explained above.

FIG. 3 shows in detail an exemplary embodiment of the transimpedanceamplifier 30 illustrated in FIG. 1. In this case, the feedback impedance30 may be formed by an impedance network in a manner corresponding toFIG. 2. In the exemplary embodiment of FIG. 3, the output signal Sa ofthe transimpedance amplifier 30 is present at the reference point 300 ofthe circuit.

The photodiode 20 serving to detect optical signals is assigned acapacitance C_(IN), which comprises the input capacitance of thephotodiode 20 and also parasitic capacitances. The capacitance C_(IN),together with the feedback impedance RF 50, forms a low-pass filterwhose limiting frequency is determined by the equationf=1/(2πC_(IN)*RF). As the impedance RF increases, the bandwidth of theamplifier is thus reduced, i.e. the maximum data rate which theamplifier 30 can amplify decreases as the value of the impedance RFincreases.

The transistors of the amplifier 30 which are explained below areembodied using bipolar technology. However, they may also be embodied asfield-effect transistors in a corresponding manner.

A first transistor T1 is present, the base (control) terminal of whichis connected to the photodiode 20. The emitter (first) terminal of thetransistor T1 is connected to ground. The collector (second) terminal ofthe transistor T1 is connected to the operating voltage U_(B), presentat a reference point 130, via a plurality of resistors R_(C1), R_(C2),R_(C3). The resistors R_(C1), R_(C2), and R_(C3) are connected inparallel with one another and together form a resistor R_(C). Two of theresistors R_(C2) R_(C3) are in this case formed such that they can beconnected in or disconnected by switches M1, M2 via correspondingcontrol terminals S1, S2. In this case, the switches M1, M2 are formedas MOS transistors, the control terminals S1, S2 being connected to therespective gate terminal. However, they may also be embodieddifferently.

Instead of or in addition to the resistors R_(C1), R_(C2), R_(C3) beingconnected in parallel in the manner illustrated, it is possible toachieve a resistor setting of the resistor R_(C) at the collectorterminal also by means of a series circuit of connectable resistorsand/or connectable capacitors.

The collector terminal of the transistor T1 is furthermore connected tothe base (control) terminal of a second transistor T2. The collector(second) terminal thereof is directly connected to the operatingvoltage. The emitter (first) terminal of the transistor T2 is connectedto ground via a mirror circuit 120 having two further transistor T3, T4.The mirror circuit 120 serves for setting a current for the transistorT2. This function may also be provided by other components, for examplean ohmic resistor.

The emitter terminal of the transistor T2 is connected to the baseterminal of the transistor T1 via the feedback impedance RF. An outputsignal Sa (cf. FIG. 1) of the amplifier 30 is tapped off at the node300.

The circuit of FIG. 3 functions as follows:

The base-emitter voltage U_(BE1) of the first transistor T1 and thebase-emitter voltage U_(BE2) of the second transistor T2 areapproximately constant. This results from the feedback to the baseterminal of the first transistors T1 via the impedance RF. Since thevoltage drop U_(RC) across the resistor R_(C) (comprising one or aplurality of the parallel-connected resistors R_(C1), R_(C2), R_(C3))depends only on the constant operating voltage U_(B) and the twobase-emitter voltages U_(BE1), U_(BE2) of T1 and T2, which are likewiseconstant to a first approximation (U_(RC)=U_(B)−U_(BE1)−U_(BE2)), theresistor R_(C) alone determines the current in the transistor T1. In theevent of supplementarily connecting or disconnecting resistors R_(C2),R_(C3) via the switches M1, M2, it is thus possible to set the currentthrough the transistor T1.

As a result, the operating point of the input transistor T1 is also setinsofar as the latter acquires, depending of R_(C), a value betweenmaximum voltage, (operating voltage U_(B)) and minimum voltage (ground).In this case, the term operating point denotes the quiescent state inthe absence of an input signal. This is described by a specific point onthe characteristic curve of the transistor.

The input signal of the photodiode 20 is amplified first by thetransistor T1 and then by the transistor T2. The current through thetransistor T2 is in this case set by the mirror circuit 120, on which areference current I_(B) is impressed.

Consideration shall now be given firstly to the case where the receiveddata have a high data rate or bandwidth. A high bandwidth is accompaniedby a low gain and a low value of RF. If a lower data rate is thenpresent, the bandwidth of the amplifier may decrease. For this purpose,it is possible, on the one hand, for the value of RF to be chosen to belarger (cf FIG. 2), which leads to a larger gain and a smallerbandwidth. Furthermore, a smaller bandwidth of the amplifier may beachieved by the resistor R_(C) between the collector terminal of thetransistor T1 and the operating voltage U_(B) being chosen to have ahigher value. This is done by disconnecting one or a plurality of theparallel-connected resistors R_(C2), R_(C3) by means of correspondingcontrol signals on the control terminals S1, S2.

The situation, then, is such that the noise of the amplifier 30 isdetermined on the one hand by the noise of the transimpedance impedanceRF and on the other hand by the noise of the transistor T1 and theoperating point thereof. In this case, it holds true that as the currentthrough the transistor T1 increases, the noise also increases. Theserelationships are described for example in H. Kressel (Editor),Semiconductor Devices for Optical Communication, 2^(nd) Edition,Springer Verlag 1982, page 89 et seq.

Disconnecting one or a plurality of the resistors R_(C2), R_(C3)increases the total resistance R_(C) between the collector terminal ofthe transistor T1 and the operating voltage. This increased totalresistance R_(C) leads to a lower current through the transistor T1 andan altered operating point of the transistor T1. Since the noise of thetransistor T1 decreases as the current through the transistor T1decreases, the noise of the amplifier 30 can be reduced in this way.

These relationships are explained in greater detail with reference toFIG. 4. In FIG. 4, the “0” curve 200 specifies the original noise curve.The illustration shows the spectral noise power as a function offrequency. Curve 200 reveals that essentially two components make acontribution to the noise of the amplifier 30. On the one hand, at lowfrequencies in the region A, a contribution to the noise is made by thenoise level of the feedback impedance R_(F). The noise of the amplifierpredominates at high frequencies in the region B. In between is atransition region C, in which the noise increases towards higherfrequencies.

In the event of a changeover of the gain of the amplifier 30 asdescribed in FIGS. 1 and 2, the noise level of the transimpedanceimpedance RF and thus the low-frequency noise component are reduced. Inconcrete terms, the value of the impedance RF is switched higher for thecase of lower data rates. The gain of the amplifier 30 is increased inthis case. At the same time, the bandwidth of the amplifier 30 isreduced, said bandwidth being approximately inversely proportional tothe gain. This is connected with the fact that, at an increased value ofRF, the limiting frequency of the low-pass filter comprising thecapacitance C_(IN) and RF decreases. Furthermore, an increase in thetransimpedance impedance RF leads to a reduced noise influence of theimpedance RF. Thus, the noise <I_(R) ²> of the impedance in thetransimpedance amplifier is defined as <I_(R) ²>=(4 kT/FR)Δf, where k isequal to Boltzmann's constant and T specifies the temperature. At aconstant temperature, the noise influence of the impedance thusdecreases as the impedance increases.

This effect is illustrated in FIG. 4. The “1” curve 210, which specifiesthe spectral noise power after a changeover of the transimpedanceimpedance RF to a higher impedance, is lowered in the region of lowfrequencies.

A further reduction of the noise is exhibited by the “2” curve 220,which specifies the spectral noise power after the changeover of thecurrent in the transistor T1 through disconnection of one or a pluralityof the resistors R_(C2), R_(C3), i.e., after increasing the resistanceR_(C) in the collector arm. Increasing the resistance R_(C) reduces thecurrent in the transistor T1. This reduction of the current in thetransistor T1 leads to a reduced noise power of the transistor T1, whichis manifested in a reduced noise power of the curve 220 at highfrequencies, wherein the noise power of the transistor predominates overthe noise of the amplifier. At the same time, the bandwidth of theamplifier is also reduced in the case of a resistance R_(C) having ahigher value. This effect also takes place, as explained, when thetransimpedance impedance RF is increased.

Consequently, an amplifier circuit 30 is described which provides anoise optimization of the amplifier on the one hand by means of achangeover of the gain of the amplifier 30 (and thus a changeover of thebandwidth of the amplifier 30) and on the other hand by means of anoperating point changeover in the input transistor T1 of the amplifier30 at lower data rates.

The configuration of the invention is not restricted to the exemplaryembodiments present above, which are to be understood merely by way ofexample. The person skilled in the art recognizes that numerousalternative embodiment variants exist which, despite their deviationfrom the exemplary embodiments described, make use of the teachingdefined in the claims below.

1. A receiver circuit comprising: an optical reception device; and anamplifier connected to the reception device, wherein the amplifierincludes: a circuit for setting the operating point of the amplifier,and also at least one control terminal of the circuit, by which anoperating point of the amplifier can be changed between at least twovalues at the user end.
 2. The receiver circuit as claimed in claim 1,wherein the amplifier further includes an input transistor, and whereinthe circuit for setting the operating point of the amplifier includesmeans for setting an operating point of the input transistor.
 3. Thereceiver circuit as claimed in claim 2, wherein the circuit for settingthe operating point of the amplifier includes means for forming asetting of the operating point of the input transistor by setting acurrent in the input transistor.
 4. The receiver circuit as claimed inclaim 3, wherein the circuit for setting the operating point of theamplifier is coupled between the input transistor and a reference point,at which the operating voltage is present.
 5. The receiver circuit asclaimed in claim 3, wherein the circuit for setting the operating pointof the amplifier comprises an impedance network with at least oneswitching device, which can be changed over by a control signal appliedto the at least one control terminal, thereby altering the totalimpedance of the impedance network.
 6. The receiver circuit as claimedin claim 5, wherein the impedance network includes a plurality of ohmicresistors which can be connected in and disconnected by means of the atleast one switching device and the at least one control terminal.
 7. Thereceiver circuit as claimed in claim 6, wherein the switching devicecomprising a switching transistor.
 8. The receiver circuit as claimed inclaim 1, wherein the amplifier further comprises at least one gaincontrol terminal, by which the gain of the amplifier can be changed overat least between two gain values at the user end.
 9. The receivercircuit as claimed in claim 8, wherein the amplifier comprises atransimpedance amplifier.
 10. The receiver circuit as claimed in claim9, wherein the amplifier includes a feedback impedance, which influencesthe gain of the amplifier.
 11. The receiver circuit as claimed in claim10, it being possible for the impedance of the feedback impedance to beset at the user end by means of the at least one gain control terminal.12. The receiver circuit as claimed in claim 11, it being possible forthe resistance of the feedback impedance to be set at the user end bymeans of the at least one gain control terminal.
 13. The receivercircuit as claimed in claim 10, wherein the feedback impedance comprisesan impedance network including at least one switching device, which canbe changed over at the user end by the at least one gain controlterminal and which alters the impedance of the feedback impedance in theevent of changeover.
 14. The receiver circuit as claimed in claim 13,wherein the switching device comprises a switching transistor.
 15. Thereceiver circuit as claimed in claim 10, wherein the feedback impedancecomprises an impedance network including at least one variableimpedance, the impedance of which can be set within a predeterminedimpedance range at least approximately linearly at the user end by meansof the gain control terminal.
 16. The receiver circuit as claimed inclaim 15, wherein the variable impedance comprises a transistor.
 17. Thereceiver circuit as claimed in claim 1, further comprising a photodiode.18. A receiver circuit comprising: an optical reception device forgenerating a data signal in response to a received optical signal; anamplifier circuit comprising: a first transistor having a controlterminal connected to receive the data signal from the optical receptiondevice, a first terminal coupled to ground, and a second terminal, asecond transistor having a control terminal connected to the secondterminal of the first transistor, a first terminal coupled to thecontrol terminal of the first transistor, and a second terminal coupledto an operating voltage source, a first configurable impedance networkconnected between the control terminal of the first transistor and thesecond terminal of the second transistor, and a second configurableimpedance network connected between the operating voltage source and thesecond terminal of the first transistor; and means for controlling thefirst and second configurable impedance networks such that, in a firstoperating mode, the first configurable impedance network generates arelatively high impedance and the second configurable impedance networkgenerates a relatively low impedance, and in a second operating mode,the first configurable impedance network generates a relatively lowimpedance and the second configurable impedance network generates arelatively high impedance.
 19. The receiver circuit according to claim18, wherein the first configurable impedance network comprises: a firstterminal connected to the second terminal of the second transistor; asecond terminal connected to the control terminal of the firsttransistor; a first impedance element connected between the first andsecond terminals; and a second impedance element and a pass transistorconnected in series between the first and second terminals, and whereinsaid controlling means comprises means for selectively turning off thepass transistor during the first operating mode, and for turning on thepass transistor during the second operating mode.
 20. The receivercircuit according to claim 18, wherein the second configurable impedancenetwork comprises: a first impedance element connected between theoperating voltage source and the second terminal of the firsttransistor; and a second impedance element and a pass transistorconnected in series between the operating voltage source and the secondterminal of the first transistor, and wherein said controlling meanscomprises means for selectively turning on the pass transistor duringthe first operating mode, and for turning off the pass transistor duringthe second operating mode.
 21. The receiver circuit according to claim18, wherein the amplifier circuit further comprises a mirror circuitconnected between the first terminal of the second transistor andground.